Autobias driving a high frequency power system

ABSTRACT

Method and apparatus are disclosed for providing a constant voltage, high frequency sinusoidal output across a varying load, using either a single or multiple switch topology operating at constant frequency while maintaining high efficiency over the entire load range. This embodiment is especially suited to applications which require the sinusoidal voltage be held very close to a desired value in the presence of rapid changes in the conductance of the load, even in the sub-microsecond time domain as is common in computer applications and the like and in powering electronics equipment, especially a distributed system and especially a system wherein low voltage at high current is required. Embodiments and sub elements provide energy storage for low voltage, high current electronic loads, an ability to supply current with rapid time variation, connection of the energy storage element to the electronic load through specially configured conductors designed to minimize the created magnetic field around said conductors, providing extremely low inductance connections, permitting larger energy storage elements to be utilized, permitting energy storage to be located relatively remotely from the powered electronic load, and a steady voltage from a transformer isolated, high frequency ac to dc converter under varying load without the necessity for feedback control, among other aspects. The addition of capacitors which interact with the leakage inductance of the transformer to produce a natural regulation condition is used and the relationship between the value of the leakage inductance of the transformer and that of the added capacitances is different from the condition of resonance at the operating frequency.

This application is a continuing application of U.S. application Ser.No. 10/222,167, filed Aug. 16, 2002, now U.S. Pat. No. 6,583,992 issuedJun. 24, 2003, which is a continuing application of U.S. applicationSer. No. 10/004,005 filed Oct. 23, 2001, now U.S. Pat. No. 6,462,964issued Oct. 8, 2002, which is a continuing application of U.S.application Ser. No. 09/534,641 filed Mar. 23, 2000, now U.S. Pat. No.6,307,757 issued Oct. 23, 2001, which claims the benefit of: U.S.application No. 60/125,768, filed on Mar. 23, 1999; U.S. application No.60/133,252, filed on May 8, 1999; U.S. application No. 60/142,102, filedon Jul. 2, 1999; and U.S. application No. 60/144,342, filed on Jul. 16,1999, all of which are hereby incorporated by reference.

BACKGROUND OF THE INVENTION

This invention specifically relates to powering computer systems whereswitch-mode DC is created to power the internal components of thesystem. It has particular applicability in new designs wheremicroprocessors have high demands and power changes. Such can relate tothe area of powering low voltage, high current electronics. Asmentioned, though, the invention is applicable in the field ofcomputing, and much of the following description is presented in thatcontext. It should be understood, however, that other embodiments are inno way limited to the field of computing, and are applicable to a widevariety of circumstances wherein a variety of power absorbing loadswhich absorb electrical power may abruptly change their power absorbingcharacteristics (that is to say, their impedance may undergo a rapidchange). They are also applicable if such loads are separated physicallysuch that the voltage which may be dropped across the dynamic impedanceof the power carrying conductors is a significant fraction of thevoltage delivered to such loads. They are also increasingly applicableto applications wherein design tradeoffs are forcing a steady decreasein operating voltages. Such situations may arise in telecommunications,radar systems, vehicle power systems and the like, as well as incomputing systems. Further, the DC/AC converter itself may haveapplications in broader and other contexts as well.

The architecture of computing systems has undergone tremendous changesin the recent past, due principally to the advance of microcomputersfrom the original four-bit chips running at hundreds of kilohertz to themost modern 32 and 64 bit microprocessors running at hundreds ofmegahertz. As the chip designers push to higher and higher speeds,problems arise which relate to thermal issues. That is, as the speed ofa circuit is increased, the internal logic switches must each dischargeits surrounding capacitance that much faster. Since the energy stored inthat capacitance is fixed (at a given voltage), as the speed isincreased, that energy, which must be dissipated in the switches, isdumped into the switch that many more times per second. Since energy persecond is defined as power, the power lost in the switches thereforeincreases directly with frequency.

On the other hand, the energy stored in a capacitance increases as thesquare of the voltage, so a capacitor charged to two volts will storeonly 44% of the energy stored in that same capacitor 10 charged to threevolts. For this reason, a microcomputer designed to operate at two voltswill, when run at the same speed, dissipate much less power than thesame microprocessor operating a three volts. So there is a tendency tolower the operating voltage of microprocessors.

Other considerations cause the microprocessor to exhibit a lower maximumspeed if operated at a lower voltage as compared to a higher operatingvoltage. That is, if a circuit is operating at full speed, and thevoltage on that circuit is simply reduced, the circuit will not operateproperly, and the speed of the circuit (the “clock speed”) may have tobe reduced. To maintain full speed capability and still operate at lowervoltage, the circuit often must be redesigned to a smaller physicalsize. Also, as the size of the circuitry is reduced, and layer thicknessis also reduced, the operating voltage may need to be lowered tomaintain adequate margin to avoid breakdown of insulating oxide layersin the devices. For the past few years, these steps have defined thecourse of microprocessor design. Key microprocessor designers, seekingthe maximum speed for their products, have therefore expendedconsiderable effort trading off the following considerations:

higher speed chips are worth more money;

higher speed chips must dissipate more heat;

there are limitations to removal of that heat;

lower voltages reduce the heat generated at a given speed; and

smaller devices run faster at a given voltage.

Of course, there are many, many important trade-off considerationsbeyond these, but the above list gives the basic elements which relateto same aspects of the current invention. The result of theseconsiderations has been for the microprocessor designers to producedesigns that operate at lower and lower voltages. Early designs operatedat five volts; this was reduced to 3.3, to 3.0, to 2.7, to 2.3, and atthe time of writing the leading designs are operating at 2.0 volts.Further reductions are in store, and it is expected that future designswill be operated at 1.8, 1.5, 1.3, 1.0, and even below one volt,eventually perhaps as low as 0.4 volts.

Meanwhile, advances in heat removal are expected to permit processors torun at higher and higher heat dissipation levels. Early chips dissipatedperhaps a watt; current designs operate at the 30 watt level, and futureheat removal designs may be able to dissipate as much as 100 watts ofpower generated by the processor. Since the power dissipated isproportional to the square of the operating voltage, even as the abilityto remove heat is improved, there remains a tendency to run at loweroperating voltages.

All of this is driven by the fundamental consideration: higher speedchips are worth more money. So the designers are driven to increase thespeed by any and all means at their disposal, and this drives the sizeof the chips smaller, the voltages lower, and the power up. As thevoltage drops the current increases for a given power, because power isvoltage times current. If at the same time improvements in heat removalpermit higher powers, the current increases still further. This meansthat the current is rising very rapidly. Early chips drew smallfractions of an ampere of supply current to operate, current designs useup to 15-50 amperes, and future designs may use as much as 100 amperesor more.

As the speed of the processors increase, the dynamics of their powersupply requirements also increase. A processor may be drawing verylittle current because it is idling, and then an event may occur (suchas the arrival of a piece of key data from a memory element or a signalfrom an outside event) which causes the processor to suddenly startrapid computation. This can produce an abrupt change in the currentdrawn by the processor, which has serious electrical consequences.

Inductance is the measure of energy storage in magnetic fields. Allcurrent-carrying conductors have associated with the current a magneticfield, which represents energy storage. It is well known by workers inthe art that the energy stored in a magnetic field is half the volumeintegral of the square of the magnetic field. Since the field islinearly related to the current in the conductor, it may be shown thatthe energy stored by a current carrying conductor is proportional tohalf the square of the current, and the constant of proportionality iscalled the “inductance” of the conductor. The energy stored in thesystem is supplied by the source of electrical current, and for a givenpower source there is a limit to the rate at which energy can besupplied, which means that the stored energy must be built up over time.Thus the presence of an energy storage mechanism naturally slows down acircuit, as the energy must be produced and metered into the magneticfield at some rate before the current can build up.

The available voltage, the inductance, and the rate of change of currentin a conductor are related by the following equation, well known bythose skilled in the art:

V=L*∂I/∂t, where L is the inductance of the conductor, and ∂I/∂t is therate of change of current in the conductor.

This equation states that the voltage required to produce a givencurrent change in a load on a power system increases as the time scaleof that change is reduced, and also increases as the inductance of anyconnection to that load is increased. As the speed of microprocessors isincreased, the time scale is reduced, and as the available voltage isreduced, this equation requires the inductance to be droppedproportionally.

Normally, in powering semiconductor devices one does not need toconsider the inductance of the connections to the device, but withmodern electronics, and especially with microprocessors, theseconsiderations force a great deal of attention to be brought to loweringthe inductance of the connections. At the current state of the art, forexample, microprocessors operate at about two volts, and can tolerate avoltage transient on their supply lines of about 7%, or 140 millivolts.These same microprocessors can require that their supply current changeat a rate of at least one-third or even nearly one ampere pernanosecond, or 3*10⁸ or 10⁹ amperes/second, respectively. The aboveequation indicates that an inductance of about 140 picohenries(1.4*10⁻¹⁰ H) and ½ nanohenry, (5*10⁻¹⁰ H) will drop a voltage of 140millivolts at these two rates of current rise. To put this number inperspective, the inductance of a wire one inch in length in free spaceis approximately 20 nanohenries, or 20,000 picohenries. While theinductance of a connection can be reduced by paralleling redundantconnections, to create a connection with an inductance of 140picohenries with conductors about a centimeter long would require some20 parallel conductors, and for instance a connection with an inductanceof ½ nanohenry would require nearly 100 parallel conductors.

The foregoing discussion implies that the source of low voltage must bephysically close to the microprocessor, which in turn implies that it bephysically small. While it might be suggested that capacitors might beused to supply energy during the delay interval required for the currentin the conductors to rise, the intrinsic inductance of the connectionsto the capacitors currently severely limits this approach. So the systemdesigner is faced with placing the source of power very close to theprocessor to ensure that the processor's power source is adequatelystable under rapid changes in current draw. This requirement will becomeincreasingly severe as the voltages drop still further and the currentsincrease, because the former reduces the allowable transient size andthe latter increases the potential rate of change of current. Bothfactors reduce the permissible inductance of the connection. This canforce the designer to use smaller capacitors which have low inductanceconnections, and because the smaller capacitors store less energy, thisdrives the power system to higher frequencies, which adds costs andlowers efficiency.

The foregoing remarks are not limited in computers to the actual centralmicroprocessor. Other elements of a modern computer, such as memorymanagement circuits, graphic display devices, high speed input outputcircuitry and other such ancillary circuitry have been increased inspeed nearly as rapidly as the central processing element, and the sameconsiderations apply.

Many modern electronics circuitry, including computers, are powered byswitchmode power conversion systems. Such a system converts incomingpower from the utility line to the voltages and currents required by theelectronic circuitry by operation of one or more switches. In low powerbusiness and consumer electronics, such as desktop personal computers,the incoming power is supplied as an alternating voltage, generally 115volts in the United States, and 220 volts in much of the rest of theworld. The frequency of alternation is either 50 or 60 Hertz, dependingupon location. Such utility power must be converted to low voltagesteady (direct) current, or dc, and regulated to a few percent in orderto be useful as power for the electronic circuits. The device whichperforms such conversion is called a “power supply”. While it ispossible to create a low voltage regulated dc power source using simpletransformers, rectifiers, and linear regulators, such units would beheavy, bulky and inefficient. In these applications it is desirable toreduce weight and size, and this approach is unsuitable for this reasonalone. In addition, the inefficiency of linear regulators is alsounacceptable. Efficiency is defined as the ratio of output power toinput power, and a low efficiency implies that heat is being developedin the unit which must be transferred to the environment to keep theunit cool. The lower the efficiency the more heat must be transferred,and this is itself a reason for finding an alternate approach.

For these reasons, virtually all modern electronics circuitry is poweredby switchmode conversion systems. These systems typically operate asfollows. The incoming utility power is first converted to unregulateddirect current by a rectifier. The rectified dc is then converted to ahigher frequency, typically hundreds of kilohertz, by electronicswitches. This higher frequency power is then transformed by a suitabletransformer to the appropriate voltage level; this transformer alsoprovides isolation from the utility power, required for safety reasons.The resulting isolated higher frequency power is then rectified again,and filtered into steady direct current for use by the electronics.Regulation of the output voltage is usually accomplished by control ofthe conduction period of the electronic switches. The resulting powerconversion unit is smaller and lighter in weight than earlier approachesbecause the size and weight of the transformer and output filter arereduced proportionally to the increase in frequency over the basicutility power frequency. All of this is well known in the prior art.

In a complex electronic system, various voltages may be required. Forexample, in a computer system the peripherals (such as disk drives) mayrequire +12 volts, some logic circuits may require +5 volts,input/output circuits may additionally require −12 volts, memoryinterface and general logic may require 3.3 volts, and the centralmicroprocessor may require 2 volts. Standards have developed so that thecentral power source (the device that is connected directly to theutility power) delivers ±12 and +5 volts, and the lower voltages arederived from the +5 supply line by additional circuitry, called voltageregulation modules, or VRMs, near to the circuits that require the lowervoltage. These additional circuits convert the +5 volt supply to highfrequency ac power again, modifying the voltage through control of theperiod of the ac power, and again re-rectifying to the lower voltage dc.

The resulting overall system is complex and not very efficient, in spiteof the use of switchmode technology. In a typical 200 watt computersystem, four watts are lost in the initial rectification of the utilityline, eight watts in the electronic switches, 2.5 watts in thetransformer, 20 watts in the output rectification and filtering, andfour watts in the connections between the center power supply and theelectronics boards. Thus 38.5 watts are lost in the conversion processfor the higher voltage electronic loads. Substantial additional lossesmay be sustained in the low voltage conversion process. A typical 50watt voltage regulation module, which may convert +5 volts at 10 amperesto +2 volts at 25 amperes for the microprocessor, will itself havelosses of about one watt each in the ac conversion and transformer, andten watts in the final rectification and filtering. Other voltageregulation modules will have losses almost as great, resulting in lossesfor the entire system which may be one-third of the power used. Someparticularly inefficient approaches may demonstrate efficiencies as lowas 50%, requiring that the input power circuits utilize twice the powerrequired by the actual final circuitry, and requiring that twice theheat be dissipated in the electronics (which must be removed by a fan)as is theoretically required by the actual operating circuitry.

This system evolved over the years and is not optimum for many currentuses, but persists because of inertia of the industry and because of theperceived benefit of maintaining industry standards on voltages andcurrents as generated by the central power unit.

An analysis of current trends in the microprocessor industry clearlyindicates that the current system will not be adequate for the future.These trends show that the current draw of critical elements such as thecore microprocessor has been steadily increasing and will continue to doso into the future. Meanwhile, the operating voltage has been steadilydecreasing, dropping with it the allowable tolerance of the supplyvoltage in absolute terms. Finally, the rate of change of processorcurrent—the current slew rate—is increasing very rapidly, withsubstantial additional increases forecast for the near future. All ofthese factors mitigate against the current technology and require a newapproach to be adopted in the future. It has been reliably estimatedthat the current powering and other technology will not last more thanone additional generation of microprocessors, and since designers arecurrently at work on the generation following the next, it can be saidthat these designers are in the process of developing a microprocessorwhich cannot be powered by currently available technology.

A further problem in the prior art is the use of square wave electronicconversion techniques. Such technology, known as PWM, for Pulse WidthModulation, produces switch voltage waveforms which have steeply risingedges. These edges produce high frequency power components which can beconducted or radiated to adjacent circuitry, interfering with theirproper operation. These high frequency power components may also beconducted or radiated to other electronic equipment such as radio ortelevision receivers, also interfering with their proper operation. Thepresence of such components requires careful design of the packaging ofthe power system to shield other circuitry from the high frequency powercomponents, and the installation of expensive and complex filters toprevent conduction of these components out of the power supply packageon its input and output leads. What is needed then, is a powerconversion system which enables small, highly efficient voltageregulation modules to be placed close to the point of power use, whichis fast overall, and which is itself efficient and at least as low incost as the prior art technology it replaces.

As mentioned earlier, the invention can be applied for use in powering awide variety of circuitry that requires low voltage and high current. Inaddition it provides capability to provide rapidly changing current. Inparticular it applies to microprocessors and similar circuitryespecially where they are requiring less than 2 volts and are projectedto require less than one volt.

SUMMARY OF THE INVENTION

It is an object of the present invention, therefore, to provide a meansfor storing energy with lower inductance connections than could beachieved with the prior art. It is a further object of the presentinvention to provide a source of energy at low voltage and high currentwhich does not need to be placed in very close proximity to theelectronic load. Similarly, it is yet an another object of the inventionto provide a source of low voltage which can sustain the voltage acrossthe powered load even in the presence of high rates of change of currentdraw

It is also an object of the present invention, to provide a means ofconverting utility power to high frequency alternating power forefficient distribution at higher efficiency than can be achieved usingexisting techniques. It is also an object to provide a means ofconverting high frequency ac power to the low dc voltages and high dccurrents required by current and future electronics at higher efficiencythan can be achieved using current techniques. It is another object ofthe present invention to maintain that efficiency over a wide range ofload conditions.

A further object of the present invention is to provide a source of highfrequency power which is substantially smaller than that of the priorart. Similarly it is an object to provide a source of low voltage athigh current which is substantially smaller than that of the prior artto permit such a source to be placed in very close proximity to theelectronic load.

It is also an object of the preset invention to provide closer controlof the output voltage of a power source, even for extremely short timeperiods. That is to say, it is an object to ease the task of thepowering or of providing a power source so that it does not need suchwide bandwidth and has a small transient response to changes in load.Thus an object is to provide a system with better transient response tochanges in load.

It is a further object of the invention to provide a power conversionsystem which stores less energy than that required by the prior art.

It is additionally an object of the present invention to provide a powerconversion system which can be produced at lower cost than the priorart.

It is also an object to address problems associated with the use ofsquare wave electronic conversion techniques. It is yet another objectof the invention to reduce possible interference between the powersystem and the electronics being powered, as well as with other devicesin the vicinity of the powered electronics, by reducing the rate of riseof currents and the rate of fall of voltages in the power system.Similarly, an object is to provide power using smoothly varyingwaveforms in the power conversion circuitry.

It yet a further object of one embodiment of the present inventionprovide to power with the aforesaid objects being satisfied, yet operateat either a constant frequency or, through other embodiments, toaccommodate variable frequencies as well.

Another fundamental aspect of the invention is the potential for theaffirmative use of the transformer leakage inductance. This can benecessary as the DC output voltage requirement is lowered.

Another benefit of this invention involves the very nature of a powersource. By incorporating some or all these elements it can be possibleto provide power remotely. By making the output capacitance consist ofonly the bypass capacitors necessary on the microprocessor pins, thecircuit feeding the microprocessor assembly can have essentially aninductive output.

Several features will be disclosed which taken together or separatelycan allow the power conversion frequency to be increased to provide alow stored energy approach to meet the high di/dt requirements for nextgeneration low voltage requirements. Thus, yet other objects includeproviding a circuit and method for providing power to electronics withlow voltage, high current and high di/dt requirements, providingsubstantially higher power conversion frequencies, providing a circuitwhich allows a reasonable amount of transformer leakage inductance andswitching device capacitances, providing a circuit or method whereby thesynchronous rectifiers (SR's) always switch with zero voltage across thedevice, allowing high frequency operation, providing a circuit or methodwhereby the control signal to the SR operates in a non-dissipativefashion, allowing HF operation, and providing a reduced size of theoutput capacitance through HF operation

Accordingly, in one embodiment the present invention is directed to asystem of energy storage which can store more energy and be placedphysically farther from the powered electronics, through the reductionof magnetic fields surrounding the electrical connections and themagnetic energy stored therein, thereby creating a faster respondingstorage and powering medium. The reduction of the magnetic fields andthe resulting reduction of inductance permits electronics to operate athigher speed, and the increased energy storage permits the poweringsystem to operate at lower speed. This reduction in powering systemfrequency may permit lower costs that could be obtained using highfrequency power systems

Similarly, the present invention in another embodiment is directed to asystem of power conversion which eliminates many of the elements of theprior art, by distributing high frequency ac power to a point near theloads, and performing a single conversion from ac to dc at the point ofpower consumption. In particular, the present invention addresses thislatter ac to dc conversion and solution of the problems related toconversion of higher voltage ac power to very low dc voltages with goodregulation and transient response.

The elimination of many of the redundant elements in the prior artapproach not only increases efficiency by eliminating a power losselement, but also reduces cost by elimination of the cost of theelements removed from the system. The reduction of frequency alsoincreases the efficiency of the powering system, because at higherfrequencies switching losses become increasingly important and may equalor exceed all other losses. The present invention accomplishes many ofthese objects by providing a low inductance connection for energystorage elements which is not limited in length through the mechanism ofreducing the volume of the magnetic field surrounding the conductorsintermediate to the energy storage element and the powered electronics.

In yet another embodiment, the present invention distributes highfrequency smoothly varying or even sinusoidal waveforms, which exhibitrelatively low rates of voltage change for a given frequency, and muchlower than alternative ac approaches, such as distribution of squarewave or trapezoidal waveforms. The distribution of sinusoidal acvoltage, rather than dc voltages as is usually done in the prior art,not only simplifies the central power unit, but also greatly simplifiesthe voltage regulation modules, reducing cost and raising efficiency.This approach also results in greatly reduced interference between thepower unit and adjacent circuitry, and simplifies the design and reducesthe cost of the line filters used to avoid conducted interference alongthe utility power lines. Also, distribution of low dc voltages (e.g., 5volts) results in relatively higher losses in the distribution wires andconnectors when compared to the use of medium voltage alternatingdistribution levels (e.g., 30 volts rms), which nevertheless remain safeto touch.

BRIEF DESCRIPTIONS OF THE DRAWINGS

FIG. 1-1 shows a conventional computer power delivery system of theprior art.

FIG. 1-2 is a more detailed depiction of a computer power deliverysystem of the prior art.

FIG. 1-3 indicates the parts of the computer power delivery system ofthe prior art that may be eliminated by the present invention.

FIG. 1-4 shows a computer power delivery system according to oneembodiment of the present invention.

FIG. 1-5 indicates an embodiment of the power conversion element of thepresent invention.

FIG. 1-6 depicts another embodiment of the power conversion element ofthe present invention.

FIG. 1-7 shows details of a switch drive according to the presentinvention.

FIG. 1-8 shows a rectifier circuit of the present invention.

FIG. 1-9 shows a variation of output voltage with changes in the valueof a capacitance in one embodiment.

FIGS. 1-10 and 1-11 show two variations of the voltage across a loadresistance as a function of the load resistance.

FIG. 1-12 shows another embodiment with a two switch configuration andvarious general elements.

FIGS. 1-13 and 1-14 are plots of voltage waveforms at various locationsfor two different loads, high and low respectively.

FIG. 2-1 shows a computer power delivery system.

FIG. 2-2 shows the magnetic field around a conductor in free space.

FIGS. 2-3a & b show the magnetic field of a confined conductor.

FIG. 2-4 shows the magnetic field configuration around a doubly confinedconductor.

FIG. 2-5 shows one embodiment of the present invention.

FIG. 2-6 shows an alternate embodiment of the present invention.

FIG. 3-1 shows a traditional buck converter of the prior art.

FIG. 3-2 shows a waveform of the center point of the buck convertershown in FIG. 3-1.

FIG. 3-3 shows an embodiment of a transformer and rectifier portionaccording to the present invention.

FIG. 3-4 shows the voltage waveforms as they may exist at variouslocations in the circuit shown in FIG. 3-3.

FIG. 3-5 shows one gate drive embodiment for the SR's according to thepresent invention.

FIG. 3-6 shows a circuit for voltage control on the primary side with asingle switching design.

FIG. 3-7 shows a family of drain to source voltages as a function of thecontrol input voltage across the FET.

FIG. 3-8 shows a circuit for voltage control on the primary side with adual switching design.

FIGS. 3-9a, b, c & d shows various synchronous rectification circuitsaccording to the invention.

FIG. 3-10 shows a bulk capacitor and a by pass capacitor arrangement asapplied to a microprocessor system in the prior art.

FIG. 3-11 shows an overall preferred embodiment of the invention using asingle switch control element.

FIG. 3-12 shows an overall preferred embodiment of the invention using adual switch control element.

FIG. 3-13 shows an overall preferred embodiment of important aspects ofthe “silver box” aspect of the design.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

As can be easily understood, the basic concepts of the present inventionmay be embodied in a variety of ways. These concepts involve bothprocesses or methods as well as devices to or which accomplish such. Inaddition, while some specific circuitry is disclosed, it should beunderstood that these not only accomplish certain methods but also canbe varied in a number of ways. Importantly, as to all of the foregoing,all of these facets should be understood to be encompassed by thisdisclosure.

In the prior art, the central power supply provides several standardvoltages for use by the electronics. Referring to FIG. 1-1, utilitypower 101, typically at 110 or 220 volt nominal ac power alternating at50 or 60 cycles, is converted by power supply 106 to standard dcvoltages, usually ±12 and +5 volts. These voltages are brought out ofthe power supply on flying leads, which form a kind of distributionsystem 107, terminated in one or more connectors 108. These standardvoltages are useful directly for powering most of the input/outputcircuitry 140 and peripherals 144, such as a hard disc, floppy disc, andcompact disc drives. As the technology of central processing unit (CPU)chip 141 has advanced, as discussed above, the operating voltage of suchchips has steadily been reduced in the quest for higher and higheroperating speeds. This increase in processor speed eventually requiredan increase in speed of the dynamic random access memory (DRAM) chips143 used to hold instructions and data for the CPU, and as a result theoperating voltage of these DRAM chips has also been reduced. Also, notall of the logic required to manage the input/output functions andparticularly the flow of data to and from the CPU and the memory andexternal devices is present on the CPU chip. These management functions,along with housekeeping functions (such as clock generation), interruptrequest handling, etc., can be dealt with by the “chip set”, shown inFIG. 1-1 as logic management circuits 145. These circuits also havesteadily increased in speed and have correspondingly required loweroperating voltages.

The standard voltages are thus too high to properly power CPU 141,memory 143, and management circuits 145. These may all require differentvoltages, as shown in FIG. 1-1, where the actual voltages shown arerepresentative only. These different voltages may each be created by anindividual Voltage Regulation Module 112 (VRM), which reduce the voltagesupplied by the power supply 106 to the voltage required by the poweredcircuitry.

From an overall point of view, the prior art process of delivering powerto a circuit load such as CPU 141 involves all of the power processinginternal to power supply 106, distribution system 107 and connectors108, and the power processing internal to VRM unit 112. This overallprocess is shown in FIG. 1-2. Central power supply 106, also called the“silver box”, uses switchmode technology, with processing elements 102,103, 104, and 105. Utility power 101 enters the silver box and isconverted to unregulated dc power by rectifier unit, or AC/DC converter102. The resulting dc power is then re-converted to alternating currentpower at a higher frequency by inverter unit 103 (also called a DC/ACconverter). The higher frequency ac is galvanically connected to and isat the voltage level of utility power 101. Safety considerations requireisolation from utility power 101, and as the required output voltage ismuch lower than that of utility power 101, voltage reduction is alsoneeded. Both of these functions are accomplished by transformer 104. Theresulting isolated, low voltage ac is then rectified to direct ormultiply direct current power output(s) by rectifier and filter unit105, distributed to the circuitry loads by distribution wiring 107 andconnectors 108. As mentioned before, specific standard voltages ±12 and+5 volts must be converted to lower voltages for CPU 141, memory 143 andmanagement logic 145, by VRM unit 112. The standard dc voltage frompower supply unit 106 (usually +5 volts) is converted to alternatingpower again by DC/AC converter 109, transformed to the lower voltage bytransformer 110, and re-rectified to the proper low voltage by AC/DCunit 111.

As the voltage of the delivered power to the circuit load is decreased,the current increases, and as the speed of CPU 141 is increased, thepower system must be able to handle larger and larger rates of change ofcurrent as well. As discussed above, this requires the source of power,which for CPU unit 141 (and other low voltage circuits) is VRM 112, tobe close to the circuit load. While for the near term designs the rateof change of current can be handled by capacitive energy storage, forfuture designs at still lower voltages and higher currents VRM unit 112must be made extraordinarily small so that it can be placed close to itscircuit load, and also must operate at a very high frequency so thatlarge amounts of energy storage are not required. The requirement forlow energy storage is rooted in the two facts that there is no physicalroom for the larger storage elements and no tolerance for their higherintrinsic inductance. Thus a requirement emerges that the frequency ofVRM 112 must be increased.

Further, a glance at FIG. 1-2 indicates at least two redundant elementswhich can be eliminated. The established policy of distributing directcurrent power requires rectifier and filter 105, and the need fordropping the voltage to lower levels requires re-conversion of the dc toalternating current power by inverter 109. One of these is clearlyredundant.

This opens the possibility of reduction of cost by eliminating elements105 and 109, and choosing to distribute alternating current powerinstead of direct current power. Of course, the AC improvement may alsobe configured with existing, traditional DC leads as well in a hybridsystem if desired. Returning to the improvement, however, as mentionedbefore, the frequency of inverter 109 has had to increase and willcontinue to increase, which requires, in the reduced system, that thefrequency of inverter 103 be increased to a level adequate to serve thefuture needs of the system. FIG. 1-3 indicates these redundant elements.

Another redundancy exists in principle, between transformers 104 and110, but the desire to provide isolated power in the distribution system107 mandates the use of transformer 104, and the requirement fordifferent voltages for the different loads may also require the variousVRMs to utilize transformer 110. Assuming that these elements are leftin place, then, the use of high frequency ac distribution produces asystem as shown in FIG. 1-4. Thus one embodiment is directedspecifically to the simplified VRM. Such an arrangement also permitselectrically remote location of power element (e.g. at locations wherethe lead inductance would have otherwise have come into play using theprior techniques).

In FIG. 1-4, central power supply 147 converts utility power 101 to dcpower by AC/DC converter 146. This dc power is then converted to highfrequency sinusoidal power by DC/AC converter 113. The sinusoidal power(or perhaps “substantially” or “approximately” sinusoidal power, as maybe produced by even a less than ideal inverter or the like) isdistributed to the location of use of the power, where high frequencyVRMs 118 convert the sinusoidal power to low voltage, high current powerfor the circuit loads such as CPU 141, input/output circuits 140, logicmanagement circuits 145, and memory 143. In this approach, a VRM isrequired not only for the aforementioned low voltage circuits, but alsofor peripherals 144, since the DC power (likely +12 volt) requirementfor these units is not supplied by the central power supply 106. (Note,the central power supply 106 may supply only sinusoidal high frequencyAC power in this approach). High Frequency Transformer 114 may thusprovide galvanic isolation and may transform the voltage from constantvoltage Sinusoidal DC/AC Converter 113 to a level considered safe totouch.

It is possible to organize a distribution system which provides aconstant current to the totality of the loads, or alternatively toprovide a constant voltage to those loads. The architecture of computersystems and other complex electronic systems with loads which requiremultiple voltages is more suited to the latter approach. That is, it isdesirable that the magnitude of the distributed ac voltage be maintainedvery close to constant against any output load variation, even on amicrosecond time scale. Thus, it can accommodate a variable load, namelya load which alters at levels which would have caused variation in thepower supplied in arrangements of the prior art. It may also beimportant to keep the Total Harmonic Distortion (THD) of the distributedac voltage low, to reduce Electro-Magnetic Interference (EMI). It shouldbe noted, however, that the present invention may be modified to providea constant current as well. That is, as those of ordinary skill in theart would readily understand, it is possible to modify the describedcircuit so that a constant current is delivered into a load which variesfrom nominal to a short circuit, for use in constant currentapplications.

Converter 113 may be designed to provide a constant output voltage withlow THD, independent of load. Some of the embodiments presented hereindepend upon being supplied with a constant input dc voltage fromconverter 146. It would of course also be possible to create thisconstant distribution voltage by feedback internal to converter 113, asan alternative, which then would not require a constant input voltagefrom converter 146. The latter approach—creating constant voltagethrough feedback—requires that the feedback system have very highbandwidth (high speed) in order to maintain the output voltage veryclose to constant against any output load variation, even on ananosecond time scale. This feedback approach may be difficult andexpensive to achieve, and the present invention is directed toaccomplishing a constant voltage from converter 113 by the intrinsicoperation of the circuit, without feedback. This can be significantbecause it can satisfy the needs of a system which has rapid energydemands such as a rapid current demand of at least about 0.2 amperes pernanosecond, at least about 0.5 amperes per nanosecond, at least about 1ampere per nanosecond, at least about 3 amperes per nanosecond, at leastabout 10 amperes er nanosecond, and even at least about 30 amperes pernanosecond and beyond. It also can be significant because it can permitreaction to a change in conditions very quickly, such as within: lessthan about a period of a “Nyquist frequency” (e.g. the Nyquist rate,that is the maximum theoretical rate at which sampling or transmissionof an event can occur for a feedback-type of system),

less than about two and a half times a period of a Nyquist frequency,

less than about five times a period of a Nyquist frequency,

less than about ten times a period of a Nyquist frequency,

less than about twice a period of said alternating power output,

less than about four times a period of said alternating power output,

less than about 200 nanoseconds,

less than about 500 nanoseconds,

less than about 1000 nanoseconds, and

less than about 2000 nanoseconds.

FIG. 1-5 shows one embodiment of a constant voltage high frequency powersource to accomplish the function of converter 113. Here dc power source119 is the circuit representation of the constant voltage from converter146, and load 128 represents the constellation of loads connected todistribution system 115 (including the effects of connectors 18 anddistribution system 115). The voltage from source 119 is converted to aconstant current by inductor 120 and either shunted by switch 122 whenthe switch is ON, or permitted to flow into network 148, comprising theelements in parallel with switch 122 when the switch is OFF. The networkthus acts as a response network, that is, one which acts after theswitch has transitioned. The voltage across switch 122 is approximatelyzero when switch 122 is ON and is dependent upon the response of network148 when switch 122 is OFF. This response waveform, or “switch voltagewaveform” is transformed by network 48 to form the voltage across load128. It turns out to be possible to choose the values of elements 123,124, 125, 126, and 127 such that the switch voltage is zero at thecommencement of the interval of time when switch 122 is ON, independentof the value of the conductance of load 128, at least within a nominalrange of conductance for load 128. This may be accomplished in thefollowing way. If the conductance of load 128 is very small (lightloading), little current will flow in inductance 127, and its value willnot strongly affect the waveform across switch 122. Then the values ofelements 123, 124, 125, and 126 may be chosen to cause the waveformacross switch 122 to be approximately zero, or to be a desired fixedvalue, at the moment when switch 122 begins to conduct. Cleardescriptions for the methodology for accomplishing this may be found inU.S. Pat. Nos. 3,919,656 and 5,187,580. Once this has been accomplished,the conductance of load 128 may be changed to the maximum nominal value,and the value of inductor 127 chosen to return the value of voltageacross switch 122 at the commencement of its ON period to the valuechosen in the first step. This algorithm will result in a circuit forwhich the value of the switch voltage waveform at the commencement ofthe ON period of switch 122 is nearly independent of the value of theconductance of load 128, within the defined nominal range. It alsoresults in a circuit for which the shape of the switch voltage waveformvaries minimally over the range of the conductance of load 128. Asignificant function of the network formed by elements 123, 124, 125,126, and 127 is to form a sinusoidal waveform across load 128. Sincethis is a linear passive network, namely, a network with no activeelements (including but not limited to steering diodes, diodesgenerally, other active elements, or the like) or a network without sometype of feedback element (an element which senses a condition and thenresponds to that condition as a result of a delayed decision-type ofresult), if the shape of the switch voltage waveform does not change inany substantial way, and especially if the fundamental frequencycomponent of the switch voltage waveform (the Fourier component of thewaveform at the operating frequency) does not change substantially, forthis circuit the value of the sinusoidal voltage across load 128 willnot change substantially. Thus selection of the values of elements 123,124, 125, 126, and 127 in this manner results in a stable, constant,high frequency, pure sinusoidal voltage across load 128, independent ofthe value of the conductance of load 128, thereby accomplishing theobjective of providing a constant voltage to the distribution system. Itshould be noted that the operation of this network to produce a constantoutput voltage is very fast; abrupt changes in the conductance of load128 anywhere over its entire nominal range may be corrected in a fewcycles of operation. This is much faster than typical feedbackapproaches could make the same correction and serves to provide a fastacting network, namely one which does not suffer the existing delay in afeedback type of system.

A unique element of the invention is its high efficiency over the entireload range from a nominal load to an open circuit or from a nominal loadto a short circuit. (As one skilled in the art should understand, oneway to achieve one as opposed to the other simply involves altering theAC distribution system by one-quarter wavelength.) This comes aboutlargely as a result of the constant switch waveform described above.Since the voltage waveform changes but little over the load range,switching losses in the circuit are not affected by load variations. Itshould also be noted that all of the benefits of this invention areobtained without changing the frequency of operation. Thus, highefficiencies such as at least about 80%, at least about 85%, at leastabout 90%, at least about 95%, at least about 98% and even at leastabout 99% efficiency and beyond can be attained.

Such a circuit, which provides a constant voltage sinusoidal outputacross a load (or even in not strictly “across” the load, moregenerically “to which the load is responsive” thus encompassing bitdirect and indirect responsiveness) which can vary at high speed,utilizing a single or multiple switch and a simple circuit, operating atconstant frequency, while maintaining high efficiency over the entireload range, is a unique aspect of this invention in the field of powerconversion.

Another unique element of the invention is in the nature of the methodof driving switch 122. As has been pointed out previously, efficiency isimportant in these applications, and it is desirable not to waste energyanywhere, including the circuit used to drive switch 122. It is in thenature of high frequency switches such as Field Effect Transistors(FETs) that they have a large input capacitance. Circuits which changethe voltage on the gate terminal in a square-wave manner must firstcharge that capacitance to a voltage well above the gate thresholdvoltage for switch 122, turning ON the FET, and in the process depositenergy into that capacitance. It must then discharge that capacitance toa voltage well below the gate threshold voltage for switch 122, in theprocess absorbing the energy stored in the gate capacitance. The powerlost in the process is the energy stored in the gate capacitance,multiplied by the frequency of operation, and this can be a substantialnumber. In the present invention this loss is avoided by affirmativelyutilizing the gate capacitance of switch 122,

thus coordinating the circuitry to the gate or capacitance. That is, theenergy stored in the gate capacitance during the period switch 122 is ONis, in the present invention, stored in another element of the systemduring the period switch 122 is OFF, and is thereby available on thenext cycle to return the gate above the threshold voltage for the nextON period. This may be accomplished by “resonating” the gate capacitance(or the effective capacitance of the system) with a series or parallelinductor. The entire system may be tuned to coordinate with thefrequency of the output and the output capacitance of the switch.Referring to FIG. 1-7, FET switch 122 is depicted as an internal switchdevice 139 with an explicit gate capacitance 138, shown separately. Gatedrive circuit 121, according to the current invention, contains inductor136 connected in series (or in parallel as shown in the dotted-linealternate connection 137) which is selected such that the reactance ofinductor 136 or 137 is equal to the reactance of capacitor 138 at thefrequency of operation. In this way the energy in the gate system istransferred from gate capacitor 138 to inductor 136 (or its alternate137) and back again each cycle, and only the inevitable losses in theinductor and gate resistance need to be regenerated for each cycle.

In such a system the gate voltage is substantially sinusoidal. It willbe obvious to those skilled in the art that the duty cycle of the system(that is, the fraction of the total period that switch 122 is ON) isdetermined by the fraction of the sinusoidal cycle which issubstantially above the threshold voltage of switch 122. It will also beobvious that, while the duty cycle of switch 122 may be controlled bythe magnitude of the sinusoidal signal, such an approach places limitson the available range of duty cycle, and also may result in longer thandesirable commutation times (that is, the fraction of the total periodduring which the switch is transitioning from the ON to the OFF state),which may increase the losses of switch 122 and thereby reduce theefficiency of the system. For this reason, the drive waveform for switch122 may be divided in the present invention into an ac portion 149 and adc portion 150, and variation in the duty cycle of switch 122 may becontrolled by varying the relative magnitude of the ac and dc componentsof the drive waveform for switch 122.

An alternative approach to constant voltage, high frequency powergeneration is shown in FIG. 1-6. Here again, dc power source 119 is thecircuit representation of the constant voltage from converter 146, andload 128 represents the constellation of loads connected to distributionsystem 115 (including the effects of connectors 108 and distributionsystem 115). Switch 122 is placed in series with inductor 129, acrosssource 119. The voltage across inductor 129 is transformed bytransformer 131 and placed across the network comprised of elements 132,133, 134, and 135. This network produces the output voltage appearingacross load 128, which again represents the constellation of loadsconnected to distribution system 115 (including the effects ofconnectors 108 and distribution system 115). Provided the values of thecircuit elements are properly chosen, this output voltage will beindependent of the value of the conductance of load 128, within anominal range of such conductance. To create this independence, it issufficient to select the values of the elements such that, as oneexample:

the reactance of inductor 129 in parallel with the magnetizinginductance of transformer 131 is equal to the reactance of capacitor 130in parallel with the adjunct output capacitance of switch 122 at thefrequency of operation;

the reactance of inductor 132 in series with the leakage inductance oftransformer 131 is equal to the reactance of capacitor 133 at thefrequency of operation; and

the reactance of inductor 134 is equal to the reactance of capacitor 35at the frequency of operation.

Selection of the values of the circuit elements in this manner willresult in a stable, constant, high frequency, pure sinusoidal voltageacross load 128, independent of the value of the conductance of load128, thereby accomplishing the objective of providing a constant voltageto the distribution system.

The necessity for the parallel resonant circuit formed by inductor 134and capacitor 135 is reduced if the minimum load conductance is not tooclose to zero. That is, the network comprised of elements 134 and 135has the function of providing a minimum load to the generator so thatthe output waveform remains sinusoidal when load 128 is removed orreduced to a very low value. Should the application to which theinvention is applied not present a load variation down to low values, orif the requirement for low THD not be present at light loads, thenetwork comprised of elements 134 and 135 may be dispensed with.Alternatively, the network comprised of elements 134 and 135 may bereduced to a single element, which may be either an inductor or acapacitor, if the highest efficiency is not demanded.

It is generally possible to also dispense with inductor 129 byaffirmatively utilizing the magnetization inductance of transformer 131.Similarly, it is generally possible to dispense with inductor 132 byaffirmatively utilizing the leakage inductance of transformer 131. Thismay be accomplished through the modification of the construction oftransformer 131 in the manner well known to those skilled in the art.

As before, to attain high efficiency it is important to affirmativelyutilize the gate capacitance of switch 122, and all the remarks madeabove in reference to FIG. 1-7 apply to the embodiment of FIG. 1-6 aswell.

As mentioned earlier, and referring to FIG. 1-4, converter 113,operating together with AC/DC element 146, is designed to provide aconstant high frequency ac output voltage with low THD, independent ofload. It is VRM 118 which must convert this high frequency ac power frompower unit 147 to low voltage high current DC power for use by thepowered circuitry 145, 141, and 143. FIG. 1-8 shows one embodiment ofthe rectifier portion of one embodiment of a VRM to accomplish thisconversion in accordance with the present invention. Input AC power frompower unit 147 may also be processed further to enhance its stabilitybefore the rectification process, and this further processing is notshown in FIG. 1-4. The result of this processing is a stable regulatedac input 177 to the rectifier circuit 178 shown in the dotted box inFIG. 1-8.

Rectifier circuit 178 is comprised of transformer 179, which in practicewill exhibit leakage inductance caused by imperfect coupling between itsprimary and secondary windings. This leakage inductance may berepresented in general as an inductance in series with the primary orsecondary of the transformer. In FIG. 1-8, it is represented by inductor180, which therefore may not be an actual component in the circuit, butrather simply a circuit representation of part of the real transformer179, as built. It should be noted that, should the natural leakageinductance of transformer 179 be smaller than desirable for any reason,additional inductance may be added in series with its secondary (orprimary) to increase the natural value, as will be understood by thoseskilled in the art. For the purposes of this disclosure, inductor 180may be considered to be the total of the natural leakage inductance oftransformer 179 and any additional discrete inductance which may havebeen added for any purpose.

Diodes 83 rectify the ac output of transformer 179, and filter inductors184 and filter capacitor 185 create a steady dc output for consumptionby the microprocessor or other electronic load 186. For small outputvoltages, the voltage drop across the diodes 183 is too large relativeto the output voltage, resulting in loss of efficiency. As a resultdiodes 183 may be profitably replaced by field effect transistor (FET)switches, which can be manufactured to have a much lower voltage drop.In this case the FET devices require a drive signal to determine theirconduction period; the circuitry to do this is not shown in FIG. 1-8.

A second problem which arises as the output voltage is dropped is theintrinsic leakage inductance of transformer 179. This inductance, which,together with other circuit inductance, is represented as inductor 180,and acts as a series impedance which increases the output impedance ofthe overall circuit. That is, there is a natural voltage divisionbetween the reactance of inductor 180 and load impedance 136, whichrequires an increased input voltage in compensation, if the outputvoltage is to remain constant over changes in the resistance of load186. This voltage division causes the output voltage to be a strongfunction of the resistance of load 186, which is another way of sayingthat the output impedance of the circuit is not small compared to theload resistance 186.

The diodes 183 shown in FIG. 1-8 would ideally conduct whenever thevoltage on their anodes was positive with respect to their cathodes, andwould not conduct when the voltage was in the opposite polarity. This iswhat is called zero voltage switching, or ZVS, because the switchingpoint, or transition, from the conducting to the nonconducting stateoccurs at zero voltage point in the waveform. Operating an FET device atZVS is an advantage, because the losses are lowered, since the devicedoes not have to discharge energy from its output capacitance, or theenergy stored in capacitors 182, which are in parallel with theswitches. As the output current through load 186 increases, the timingfor the switches to produce ZVS must change, and may complicate the FETdrive circuitry. In the description of the figures which follow, weshall nevertheless assume that the switches are operated at ZVSconditions, or that a true diode is used.

FIG. 1-9 shows how the output voltage varies with changes in the valueof capacitance 182 placed across diodes 183. These curves were plottedfor an operating frequency of 3.39 MHz. As may be seen in FIG. 1-9, asthe value of capacitances 182 are increased, the output voltage (thatis, the voltage across load resistance 186) first begins to increase,but as the value of the capacitance 182 is increased still further, thevoltage across load resistance 186 begins to drop again. Thus there isan optimum value for the capacitances 182 which obtains the highestvoltage transfer function. In FIG. 1-9 two curves are shown, curve 187 avalue of inductance 180 of 40 nH, and curve 188 for a value ofinductance 180 of 20 nH. Curve 187 shows that a peak in output voltageoccurs at a value for capacitances 182 of about 27 nF, while curve 188shows that a peak occurs at a value for capacitances 182 of about 86 nF.Note that these are not a factor of two apart (86/27>3) as would be thecase if the values of capacitances 182 and inductor 180 satisfied theresonance condition since the two curves are for values of inductor 180which are a factor of two apart. This means that the condition formaximum output is not the same as for resonance at the frequency of theinput power from generator 177. The two capacitors 180 may be replacedby a single capacitor 181 in a parallel position across the secondarywinding of transformer 179 and inductor 180, with the same result,although the current in the diodes 183 will not be the same in thiscase.

FIGS. 1-10 and 1-11 show the voltage across load resistance 186 as afunction of the load resistance 186. The slope of these curves is ameasure of the output impedance of the circuit 178. That is, if theslope is zero, the output impedance is zero, and the circuit exhibits“natural regulation” without feedback. Curve 189 in FIG. 1-10 and curve192 in FIG. 1-11 show that, for a value for capacitances 182 equal tothe value which results in a peak in voltage across load resistance 186,a slope of nearly zero is obtained, without feedback. That is, for aproper selection of the value of capacitances 182 in relation toinductance 180, the voltage across load resistor 186 becomes relativelyindependent of the actual value of the load resistor 186— the output is“naturally regulated”. It will be seen that the advantage of “naturalregulation”—regulation without feedback—is that one does not need towait for a feedback system to recognize a change in output voltagecompared to a reference, and to change some parameter internal to thecircuit. Under the described conditions, the output voltage is heldconstant and maintained so within a cycle or two of the operatingfrequency, which is short compared to stable feedback systems.

Thus a system has been described which produces a stable output voltageover a wide range of load resistances without feedback, even underconditions of rapid change in the load resistance. For systems which cantolerate the change in output shown in the figures, no feedback isrequired. For systems which require tighter control of the outputvoltage under conditions of changing load, feedback may be added, and itwill be noted that the teachings of the present invention reduce therequirement for action on the part of the feedback system, permittingsimpler, faster, and less costly feedback circuits to be used.

As mentioned earlier, the circuit can be embodied in a variety ofmanners to achieve the overall goals of this invention. For example,referring to FIG. 1-12 as but one other example of a circuit design, ingeneral, the circuit can be understood. It may have any combination of avariety more generically stated elements. First, it may have a constantoutput element, such as the constant output voltage element 161. In thisarrangement, the constant output element serves to maintain some outputparameter as a constant regardless of a variation such as may occur fromthe variable load. As one skilled in the art would readily understand,the parameter maintained may be selected from a great variety ofparameters, including but not limited to parameters such as:

a substantially constant switch voltage output which is substantiallyconstant over all levels at which said variable load exists practically,

a substantially constant load voltage input which is substantiallyconstant over all levels at which said variable load exists practically,

a substantially constant switch voltage Fourier transform which issubstantially constant over all levels at which said variable loadexists practically,

a substantially constant switch voltage output waveform which issubstantially constant over all levels at which said variable loadexists practically,

a substantially constant switch voltage transition endpoint which issubstantially constant over all levels at which said variable loadexists practically, and

all permutations and combinations of each of the above

In the configuration shown, this constant output voltage element 161 hasinductor L1 and capacitor C5 which may be tuned for series resonance atthe fundamental frequency of operation, inductor L2 and capacitor C6which may be tuned for parallel resonance at the fundamental frequencyof operation, and capacitors C7 & C8 arranged to form a half supply withlow AC impedance as is common for a half bridge configuration, with R5representing the load to be powered. Of course, from these generalprinciples, as a person of ordinary skill in the art would readilyunderstand, other designs can be configured to achieve this basic goal.

Second, the system can include a constant trajectory element such as theconstant trajectory element 162. In this arrangement, the constanttrajectory element serves to maintain the response waveform (or even theFourier component of the waveform) as substantially a constantregardless of a variation such as may occur from the variable load. Inthe configuration shown, this constant trajectory element 162 hasinductor L4 connected to a half supply (shown as capacitors C7 & C8). Itprovides a constant current at the time of transition from switch T1conducting or switch T2 conducting (or visa versa) where diode D2 andcapacitor C2 are adjunct elements of switch T1, and diode D3 andcapacitor C4 are adjunct elements of switch T2. The trajectory which ismaintained may even be held to one which present a continuous secondderivative of voltage with respect to time. As shown herein, designs mayalso be configured to achieve a constant end point. The end point may ormay not be zero, for instance, it may be desirable in certain designs tohave a non-zero end point. That type of a design may include values suchas: zero volts, a voltage which is less than a diode turn-on level, lessthan about 5% of said switch DC supply voltage, less than about 10% ofsaid switch DC supply voltage, less than about 20% of said switch DCsupply voltage, and less than about 50% of said switch DC supplyvoltage, each over all levels at which said variable load existspractically. Regardless, a constant result (trajectory, end point, orotherwise) can be important since it is the voltage at the moment ofswitch turn-on or the avoidance of turning on the body diode which canbe highly important. Again, from all these general principles, as aperson of ordinary skill in the art would readily understand, otherdesigns can be configured to achieve each of these basic goals. Designsmay thus provide a network which is substantially load independent andwhich provides a substantially trajectory fixed response. Further, anynonlinear transfer characteristics of any component, such as thevaractor capacitance nature of many switches, the nonlinear transfercharacteristics of a transformer, or the like, can be affirmativelyutilized by the network as well for an optimal result.

Third, the circuit may include an energy maintenance element, such asthe energy maintenance circuit 163. In this feature, the energymaintenance circuit 163 serves to maintain the energy needed as aconstant regardless of a variation such as may occur from the variableload. In the configuration shown, this energy maintenance circuit 163has a capacitor C6 configured in parallel with inductor L2, both beingin parallel with the load shown as R5. This element may serve to supplysubstantially all of the rapid energy demand of the load such asdiscussed earlier. Again, as before other designs can be configured toachieve this basic goal.

Fourth, the circuit may have some type of stabilizer element such as thestabilizer element 164 shown. This stabilizer element 164 serves toabsorb energy not in the fundamental frequency in accordance with theprinciples discussed in U.S. Pat. No. 5,747,935, hereby incorporated byreference, to the assignee of the present invention.

Finally, the circuit may include an automatic bias network such as thedirect bias alteration element 165 as shown for each switch. In thisarrangement, these networks may include some type of voltage divider 166with a conduction control element such as diode 167. Here, the voltagedivider 166 uses two resistors R1 & R2 which may be selected to beequal, each with high values such as 1 k ohm. This element provides anegative bias in proportion to the AC drive amplitude. The result can bea conduction period which is independent of the drive amplitude. It canthus provide a constant dead time (response time) when neither switch isin the conductive state. Again, from these general principles, as aperson of ordinary skill in the art would readily understand, otherdesigns can be configured to achieve this basic goal as well.

As illustrated in FIGS. 1-13 and 1-14, it can be seen how a properlyconfigured system according to the present invention has the constancyfeatures mentioned. The plots 1-4 show waveforms as follows:

1—the voltage at the junction between switches T1 and T2;

2—the output voltage across the load, R5;

3—the current through L1; and

4—the current through L4.

By comparing the high load and low load situations for the same networkas shown between the two figures, several events can be noticed. Theseinclude the constant output voltage (A), constant end point (B and B′),constant trajectory (C and C′), constant response time period (D andD′), zero voltage switching (B and B′), and constantly an event of zeroload current in the transition (E), all even though there is a highlyvarying power and load current as indicated by the current into thenetwork at L1 (F and F′). Other features are also noticeable, as oneskilled in the art should easily understand.

Further, in yet another embodiment aspects such as feed forward andcounteracting any low frequency ripple superimposed on the dc inputvoltage can be addressed. For instance, a modulation of the outputvoltage waveform by said low frequency ripple can be counteracted bysuperposition of a signal derived from said low frequency ripple on theinput drive signal.

FIGS. 2-1 to 2-6 shows yet another aspect of the invention. Referring toFIG. 2-1, powering source 201 is connected by conductors 202, which maybe wires or alternatively perhaps by traces on a printed wiring board,to the powered electronics 204. The wires or traces 202 exhibit anintrinsic connection inductance 203, which schematically may beconsidered in series with the source 201. In the absence of energystorage element 207, should the current 209 drawn by the poweredelectronics 4 change abruptly, the presence of inductance 203 causes adrop δV in voltage 208 across powered electronics 204 in accordance withthe formula

V=L*∂I/∂t, where L is the value of inductance 3, and ∂I/∂t is the rateof change of current in conductors 2.

This drop may cause improper operation of electronics 204, and istherefore undesirable. To reduce the magnitude of the drop, energystorage element 207 is conventionally placed close to electronics 204,through wires or circuit traces 205. This element is placed close toelectronics 204 so that the intrinsic inductance 206 of the wiring 205is small, and so does not create a significant voltage drop betweenenergy storage element 207 and powered electronics 204. Thus the powerrequired by the powered electronics 204 is supplied by energy storageelement 207 during the short period immediately following any change incurrent requirement by electronics 204, preventing the surge in currentfrom flowing through conductors 202, and thus preventing the drop. Putin other words, if the inductance 206 is small compared with inductance203, most of the change in current 209 drawn by electronics 204 occursin circuit loop 210 containing energy storage element 207, leaving thecurrent in inductances 203 constant. This has the effect of producing arelatively constant voltage 208 across electronics 204.

Heretofore, in order to provide this effect, the conductors 205 had tobe kept very short in order to keep their intrinsic inductance 206 lowenough to keep voltage 208 constant. This is so because the longer theconductors the larger the volume of magnetic field created by thecurrent through them. The energy stored in the magnetic field isrepresented by the volume integral${E = {\frac{\mu_{0}}{2}{\int_{\tau}^{\quad}{H^{2}\quad {\tau}}}}},$

where μ₀ is the permeability of free space, H is the magnetic field, andthe integration is carried out throughout the space around theconductor.

The energy in an electrical circuit carrying current may also berepresented by the equation E=½L I². Since the magnetic field H around aconductor is proportional to I, the current through it, L isproportional to the volume, and it will be seen that if the magneticfield can be confined to smaller volume the conductor will exhibit lessinductance.

FIG. 2-2 shows a current carrying conductor 211 in free space, with themagnetic field 213 around it created by the current 212 through it.Magnetic field 213 exists throughout all space outside of the wire, withthe field at a particular radius shown as representative field line 213.As the field falls off with distance from the wire only linearly withthe distance from the wire, the field generally fills a large volume andsignificant energy is stored.

FIG. 2-3a shows a confined current carrying conductor 211, by which wemean a current carrying conductor with another conductor 214 placednearby. If the nearby conductor 214 carries an equal but oppositecurrent 215 to the principal conductor 211, the fields will be largelyconfined to the space between the two conductors. FIG. 2-3b shows across-section of the conductors, in which is shown that the field 217from conductor 211 cancels the field 218 from conductor 214 outside ofthe conductors, but reinforces the field from conductor 214 in the spacebetween the two conductors. This produces the resulting field 216between the two conductors. The smaller the distance d between the twoconductors, the more confined the field, and the smaller the inductanceof the conductor system 211 and 214.

The cancellation of the field outside the conductors is obtained becausethe field due to each of the two conductors 211 and 214 are identical,except for the slight offset in physical location of the two conductors,provided the current is the same in the two conductors. This may beobtained directly, or by induction. To understand the latter, note thatif the current in conductor 211 changes rapidly, the field 217 will alsovary rapidly, which will induce a current in conductor 214 placed nearbythrough the principle of induction. This current 215 will naturally besuch to oppose the field 217 in the space beyond it, and thereby willreduce volume occupied by the field 217. This will reduce the inductanceof conductor 211. In this case it is important not to connect the endsof conductor 214 to any part of the circuit of conductor 211; thecurrent in conductor 214 must in this case be obtained entirely byinduction. As shown in FIG. 2-4, this principle may be extended by theplacement of a second external conductor 219 with induced current 220 onthe other side of conductor 211, and it will be obvious that this may befurther extended by placement of further conductors in other nearbyplanes at other angles to conductor 211.

The instant invention applies this general principle to the problem ofproviding energy storage for electronic loads whose current draw mayvary rapidly. One embodiment of the invention is shown in FIG. 2-5. Hereelectronic load 204, which may be a microprocessor or any other loadwhose current may vary rapidly, is situated on printed wiring board 222.Energy storage element 7, which is situated a distance d from load 204,is connected thereto by a pair of conducting traces 221. As previouslyindicated, it is important that these traces be as close to one anotheras possible, and therefore may be preferably implemented as two interiortraces on a multi layer printed wiring board with very thin insulationtherebetween. In this embodiment, the currents are directly forced to beequal in the conductor pair 221 by the expedient of making them the twoconductors indicated as 205 in FIG. 2-1. Since these two conductors arepart of loop 210, their currents are equal and opposite, and theirmagnetic fields largely cancel, lowering their inductance considerably.

FIG. 2-6 shows another embodiment of the invention. Here electronic load204 is located as before on printed wiring board 222. Energy storageelement 207, however, is located away from board 222 on its own separateprinted wiring board 223. Connection between electronic load 4 andenergy storage element 207 is made with a flexible printed wiring board225 here used as a cable; this cable consists of two parallel planes ofconducting material on a flexible substrate, with the planes separatedby as little distance as possible (i.e., the substrate should be as thinas possible). Flexible wiring board 225 is in turn connected toelectronic load 204 and energy storage element 207 by connectors 224,whose purpose here is to permit easy assembly and disassembly of theensemble (and therefore may of course be dispensed with if desired).Here connectors 224 are located very close to load 204 and element 207to prevent the inductance of the connections therebetween from being alimiting factor. It will be obvious to those skilled in the art that theembodiment of FIG. 2-5 may be combined with that of FIG. 2-6, withconductor elements such as the conductors 221 shown in FIG. 2-5providing a connection between connectors 224 and load 204 and element207 in FIG. 2-6, permitting connectors 224 to be located farther fromload 204 or element 207 as desired, or as may be made necessary by otherprinted wiring board layout considerations.

It may be necessary to further reduce the inductance of the connectionsto the energy storage system, and this may be accomplished by placing aplurality of conductor systems such as herein described in parallel.

Alternatively, there may be more than one energy storage element in thesystem, and these may be connected in parallel across the powered load204 with a plurality of low inductance connections 5 of the type hereindescribed. Such systems with a plurality of energy storage elements areexpressly envisioned as part of the instant invention. Also,conventional means for connection of energy storage elements may becombined with the instant invention to provide a hybrid approach for theconvenience of the designer.

Through these various embodiments, and others that will be apparent tothose skilled in the art, it is possible to connect an energy storageelement 207 relatively remotely from the powered electronic load 204,which may thereby exhibit rapid changes in load current without creatingtransient voltages on its powered leads. Thus the invention enablesdesign tradeoffs heretofore unavailable to the designer. For example,there may be limited space adjacent to the powered load 204, and in theprior art this limited the amount of energy storage the designer mightplace there. This in turn either forced the design of load 204 so thatit exhibited lower rates of change of current draw, or forced the use ofhigher frequencies in the design of powering source 201, so that theenergy in the limited energy storage might be replenished more often.The present invention places the energy storage element remote to thepowered load 204, perhaps even on a different circuit board, where muchmore space might be available. Thus the energy storage element may bemade bigger (i.e., capable of storing more energy) and this in turn canbe used to permit larger rates of change of current draw in powered load204, or alternatively use of a lower frequency design for poweringsource 201, or a combination thereof.

Buck converter topologies shown in FIG. 3-1 are in current use forpowering microprocessors. For a 2.5 volt, 13 ampere requirement, aswitching frequency of 300 kHz is becoming inadequate. To meetsubstantial step load changes an output capacitance 301 of 3 mF(millifarads) is becoming required. As microprocessor voltagerequirements move downward toward 1.0 volt at 50 amperes, the prior arttopologies become even less suitable. With a drop in voltage (and anattendant drop in differential voltage tolerance) of 2.5 times, and anincrease of current of 4 times, an output capacitor of 30 mF would beneeded to maintain the required step response. It becomes increasinglydifficult or impossible, however, to locate such a large capacitor closeto the microprocessor connections. In addition, the cost of thisapproach increases with decreasing voltage. The other possibility wouldbe to increase the frequency. The voltage waveform 302 shown in FIG. 3-2is typical for a buck converter. When the frequency increases in such anarrangement, however, the non-resonant edges of this waveform causeproblems such as the commutation of FET output capacitance and preventincreasing the switching frequency above about a megahertz. Thissituation is rapidly becoming serious as microprocessors and other lowvoltage electronics are being developed which are increasingly difficultto provide suitable power for. The present invention permits theachievement of higher frequencies and currents as will be required. Itpermits frequencies such as greater than at least about 300 kHz, greaterthan at least about 500 kHz, greater than at least about 1 MHz, greaterthan at least about 3 MHz, greater than at least about 10 MHz, and evengreater than at least about 30 MHz and beyond, and can be configured tohandle currents of more than about 15 amperes, more than about 20amperes, more than about 50 amperes, and even more than about 100amperes and beyond.

In one embodiment, an aspect of this invention is the basic change froma circuit converting dc to dc to a circuit transforming ac to dc makinguse of a transformer and a synchronous rectifier. A transformer isuseful in this approach as it is possible to eliminate large currentsbeing distributed to the converter input. The high current secondary canthus be located physically close to the load. One circuit foraccomplishing this shown in FIG. 3-3.

With the invention disclosed, the energy conversion frequency can beincreased substantially, thereby allowing the output capacitance 303 toremain small and be located adjacent to a given load such as themicroprocessor interconnections. In fact, much higher conversionfrequencies can be achieved and whereby the output capacitance can besubstantially reduced. In the case of the 1.0 volt, 50 ampererequirement, the output capacitance 303 with the present invention canbe 500 μF or lower, depending upon load requirements. In fact, with thepresent invention, designs can be accomplished which provide a networkhaving an effective capacitance (that which causes an appreciable effectin the use or circuit designed) which is less than about 10 millifarads,less than about 3 millifarads, less than about 1 millifarads, less thanabout 0.5 millifarads, and even less than about 0.3 millifarads.

Such a dramatic improvement can come through the incorporation ofseveral elements individually or simultaneously. One primary goal ofthis invention is the elimination of frequency related limitations.Consequently it can be important to eliminate forced voltage commutationof any capacitors. The Synchronous Rectifier (SR) 304 device used may bea Field Effect Transistor (FET) with adjunct drain to source capacitance305. This SR can always be commutated to the conducting state at a timewhen there is zero voltage across it.

FIG. 3-3 shows a preferred embodiment for the rectification portion of alow voltage high current supply. The element LT 306 (total seriesinductance) is defined as the total of the transformer leakageinductance plus any other inductance in series with the transformer(inductance in the primary is simply scaled to the secondary). Theelement CT (total parallel capacitance) is defined as the total of theSR adjunct capacitance 305 (Coss), plus any external parallelcapacitance of each SR 307 (Csr) plus any capacitance in parallel withthe transformer secondary 308 (Cp).

There are several parameters which may be considered to optimize thiscircuit. If the load being powered has the possibility of high di/dt orif the load current can be a step function up or down then the followingparameters could be considered:

fundamental frequency of operation

transformer turns ratio

LT

CT

conduction angle (CA) for the SR's

phase delay (PD) of the SR's

The output inductance LF and capacitance CF can be important but mayhave a less direct impact on the proper operation of the invention.

Also to be potentially considered is the basic relationship betweenconduction angle and efficiency. In prior art and practice theconduction angle for the SR's has been carefully chosen to be less thanor equal to 180 degrees (i.e., no SR conduction overlap) to prevent ashort circuit on the transformer secondary. This common misperceptionarises from lower frequency assumptions. With the present invention, aconduction angle greater than 180 degrees is not only allowed butprovides a fundamental benefit of operation. Conduction angles in therange of 300 degrees or higher are clearly demonstrated. With properlychosen LT, CT, phase angle (PA) and conduction angle (CA), the drainwaveforms on the SR's 304 shown in FIG. 3-4 can be realized. With theseconditions, a low ratio of SR root-mean-square (RMS) current to outputcurrent can be realized. Ratios of <1.3:1 have been achieved.

Just as a general comparison, the waveforms from FIG. 3-4 can becompared to FIG. 3-2 from the prior art. They both share the low dutycycle aspect but it is clear in FIG. 3-4 the switching of the SR occursat zero volts and is ideally lossless.

Leakage Inductance & Overlapping Conduction Angle:

The transformer leakage inductance is a fundamental limiting factor forlow voltage, high current, high frequency power supplies. It consists ofan inductance in series with the transformer and has historicallylimited the conversion frequency.

In prior art leakage inductance has been dealt with in various ways.Three patents, by Schlecht, Lee and Bowman, covering dc to dc converterswill be touched on as all include methods of handling the leakageinductance. In Schlecht et al., U.S. Pat. No. 4,788,634, the leakageinductance is managed by minimizing it. As that patent states: “It isdesirable to limit the size of this leakage inductance to a negligiblysmall value compared to the resonant inductor [in this case thetransformer primary inductance] such that the unilateral conductingelement and controllable switch both have zero voltage switchingtransitions.” In Lee et al., U.S. Pat. No. 4,785,387 and Bowman U.S.Pat. No. 4,605,999 the transformer leakage inductance is used in acircuit resonant at or slightly above the fundamental frequency. Thegoal for this circuit is to accomplish zero voltage switching both forthe primary switches as well as for the rectifiers. However, the presentinvention shows use of the leakage inductance in a manner not resonantat the fundamental frequency.

One fundamental aspect of this invention is a circuit topology and classof operation which can make allowance for a larger leakage inductance.This benefit can be realized by the choice of a high conduction angle inthe SR's. In fact, for some applications conduction angles even greaterthan 300 degrees are shown to be valuable. As the output voltagerequirement is reduced and the current requirement is increased, both ofthese shifts result in still higher conduction angles. The setting ofthis large conduction angle, the total inductance and total capacitanceis done simultaneously with one of the desirable conditions being ZeroVoltage Switching (ZVS) for the synchronous rectifiers. This allowsoperation at a higher frequency or, at a given frequency operation witha higher leakage inductance. This combination of high frequencyoperation and/or higher leakage inductance tolerance is a fundamentalbenefit of this design and may perhaps be a necessary benefit asmicroprocessor power requirements become more difficult to fulfill.

One additional note with respect to the total capacitance—the choice oflocation between putting the capacitor across the transformer 308 oracross the SR's 307 changes the current waveform through the SR's butdoes not greatly affect the voltage waveform. With the capacitor acrossthe transformer makes the current waveform more like a square wave whileit is quasi-sinusoidal when the capacitor is across the SR. Thisdifference can have significant ramifications as those of ordinary skillin the art should readily understand to some degree.

High Voltage on SR:

One general principal observed in rectifier circuit design is tominimize the reverse voltage stress across the rectifier device.Depending on the type of filter input the peak inverse voltage isusually in the range of being equal to the DC output voltage upwards to1.4 times the output voltage or in rare circumstances up to twice the DCoutput voltage.

One consequence though of the high conduction angle is substantiallyhigher voltage across the rectifier devices. For example in the circuitvalues disclosed here the output voltage is 1.8 volts while the voltageacross the rectifier devices is 15 volts! Historically this type ofcircuit performance has been thought of as poor practice for a varietyof reasons as those of ordinary skill in the art well understand.Perhaps this is one reason why such a valuable circuit has not beendiscovered to date.

But a high conductance angle with attendant high voltage across the SRduring the non-conducting state has the benefit of low RMS currentthrough the SR during the conducting state and is a condition forallowing large transformer leakage inductance. This circuit is ideallysuited for low voltage, high current requirements. Furthermore it iswell suited to loads which have a high di/dt requirement as a result ofthe higher operating frequency and lower stored energy in the outputcapacitance. As it turns out the higher voltage requirement for the SR'sis not troublesome. With current manufacturing technology there appearsto little benefit to restraining the SR off state voltage to less thanabout 20 volts.

Gate Drive:

The next circuit being disclosed, FIG. 3-5, is a gate drive circuit thatderives its power from the ac input and uses only passive elements. Thegate drive of the SR's is also almost lossless. This all results in lowcost and predictable performance. It is also important for higherfrequency operation.

In addition it is possible to add a dc or low frequency bias to provideregulation or improve efficiency under various load conditions. In FIG.3-5 the point labeled BIAS INPUT is an example of an injection point forthe control input. Varying the voltage on this input has the effect ofvarying the conduction angle of the SR's without effecting the DELAYANGLE (FIG. 3-4).

The correct phase angle for conduction of the SR's is determined by thegate drive. Referring to FIG. 3-4, the angle labeled DELAY ANGLE couldbe derived by using something like elements L1, R1,2 and C1,2 of FIG.3-5. The inductance L1 includes the gate drive transformer leakageinductance.

There could be many variations of gate drive which embody theseprinciples. This may be contrasted with conventional technology in whichthe gate drive is derived from a DC source and involves timing circuitryand switching devices.

Regulation with the SR:

It is possible to also control and/or regulate the output voltage byvarying the SR Conduction Angle (CA). Consider FIG. 3-3 again with theinclusion of the capacitor Cin 309 shown in dotted lines.

To select values for the controlled output circuit, first examine thecase where the CA goes to 360 degrees for the SR's. This results in azero DC output. The impedance of Cin 309 should now be matched to thevalue of LT (transformed to the primary by the square of the turnsratio) forming a parallel resonant circuit at the fundamental frequency.As can now be seen the AC input is only loaded by a parallel resonantcircuit which in the ideal condition is lossless.

There exists a continuum of CA's from 360 degrees downward until thefull load condition is reached as before. With properly chosen circuitparameters ZVS switching can be maintained over the whole regulationrange. One important requirement for ZVS is to provide constant phaserelationship between conduction time and the ac input. In the firstorder analysis, the only control input required is that shown in FIG.3-5.

Parametric Regulation

Another method of providing regulation or control of the output could beto use parametric elements such as a varactor capacitor or saturableinductor to vary the output voltage. This can involve tuning the circuitto maximize the sensitivity to a given element and subsequently varyingit. Another approach to this type of design is to begin with a basictransfer function having the characteristic of a voltage source. Thenwith small changes in one or more variable elements, the output can beheld constant.

For some load requirements, this method of control may be the simplestor most cost effective. In particular loads which do not have high di/dtrequirements or if the voltage required is not too low, parametricregulation may be ideal.

This method of control may have the disadvantage of poor response timefor varying loads and poor input regulation. Another disadvantage is theincumbent increased sensitivity to component tolerances. In FIG. 3-4 itcan be seen that the CA is quite large. In general, the optimum CAincreases for lower output voltages. One consequence when usingparametric regulation is that it can become increasingly difficult tomanage the increased sensitivity of the output voltage to the actualcircuit values. If the component sensitivity becomes unmanageable, itmay be preferable to optimize the rectification portion of the circuitfor rectification only and regulate or control on the primary side ofthe transformer, where the impedance is higher. Layout and componentvalues can be more manageable on the primary.

Regulation on the Primary Side (with a Single Ended Switch):

FIG. 3-6 shows a simplified series switch on the primary side of thetransformer. This circuit design can be used to vary the ac voltage onthe input of the transformer as a potential method of regulating the dcoutput. For instance, C1 310 can be resonant with any residual inductivecomponent of the rectifier circuit. C2 311 may be low impedance at thefundamental frequency. The duty cycle of Q1 312 can be controlled tovary the ac voltage into the rectifier circuit. The phase delay 313 (L1,R1, and C4) may be chosen such that at the commencement of conductionthe voltage across Q1 312 is substantially zero. Further, the gate driveof Q1 can be set in similar fashion to the gate drive for thesynchronous rectifier discussed earlier. The ac input 315 may be used asthe source power, transformed down in voltage and supplied to the gatethrough the delay circuit 313. In series with this drive signal can be acontrol input 314. By summing these two voltages the conduction anglecan be varied from 0 to 360 degrees.

The conduction angle can be set by the control input and the phaserelationship may be derived from the ac input 315. With properly chosencircuit elements and delay time, Q1 312 may be always commutated to theconducting state at a time when the voltage across it is zero. Thus theac voltage to the rectifier circuit can be varied from nearly zero tofull while maintaining a lossless condition. FIG. 3-7 shows a family ofvoltage waveforms across Q1 312 (Vds for a FET switch) as a function ofthe control input. The waveform labeled 316 occurs with a low bias thatresults in a short conduction time. This condition provides minimumoutput. The waveform labeled 320 occurs with a high bias input andcorresponds to a large conduction angle and provides maximum output. Asimultaneous optimization of all parameters is also possible.

Regulation on the Primary Side (with a Dual Switch):

FIGS. 3-8 and 3-12 show other arrangements to provide regulation on theprimary side of the transformer. This circuit can use two switches 323that may operate 180 degrees out of phase. They can operate so as tomove from a series resonance between a capacitor 321 and the leakageinductance 322 of the series transformer 320. This occurs when bothswitches are closed. This shorts the primary inductance and leaves onlya series resonance already mentioned. This condition can give maximum acvoltage to the rectifier circuit.

A second condition can occur when both switches are completely open.During this condition the capacitors 324 (which includes the switchadjunct capacitance) can be in series across the series switchtransformer. It is also possible to just use a capacitor across thetransformer 325 or a combination of both. This total capacitance can beresonant with the magnetizing inductance of the transformer. This cancreate a parallel resonant circuit in series with the primary of themain transformer and may result in minimum ac voltage to the rectifiercircuit.

The third and normal condition can occur with a variable conductionangle. With the values disclosed this circuit can operate over theentire conduction range with ZVS.

Natural Regulation

If certain values of total inductance, total capacitance and the outputfilter inductance are chosen correctly a new phenomenon can exist. TheDC output voltage can remain relatively independent of the load current.This can occur without any variable elements or feedback.

EXAMPLES

Choosing all the circuit parametric values can be a lengthy task. Thefollowing example is a general-purpose rectifier which may be optimizedfor powering a microprocessor operating at 1.8 volts and requiring 20amperes. Using the circuit of FIG. 3-3 the following parametric valuesmay be appropriate:

Frequency=3.3 MHz

Turns ratio=5:1

Input voltage=30 VAC

LT=30 nH

CT=10 nF

Cin=2 nF

L1 & L2=100 nH

Co=500 μF

SR1 & SR2=3 ea. FDS6880

Conduction angle=266 degrees

Delay angle=24 degrees

FIG. 3-5 shows one embodiment of a SR gate drive; it consists ofsummation of sinusoidal signal derived from the AC input plus a controlsignal. Also, the signal derived from the AC input can have an optimaldelay for high efficiency. This circuit can produce a clean ac voltageby taking advantage of the gate transformer leakage inductance and thegate capacitance to filter harmonics from the ac input. This circuit canalso show the creation of delay using R1,2, the combination of C1,2(which includes the adjunct gate to source capacitance), and theinductor L1.

Output Trap

Also shown in FIG. 5 is a valuable filter element. C3 and L1 can form aparallel circuit resonant at twice the fundamental frequency. Thisparallel trap can provide the following advantages:

1) targeting largest ripple component only

2) storing very little energy—allowing fast loop control

3) sharply reducing the ac current component of the connection to theoutput capacitor. If this circuit powers a microprocessor, the C4 may becritically located to minimize inductance to the microprocessor. In thiscase the parallel trap can minimize the ‘hot leads’ problem for theconnection from the rest of the circuit to the Cout.

Topology Variations

FIG. 3-9 A, B, C, and D show various topologies that may be used toimplement the invention disclosed. The location of the total inductanceand total capacitance is shown in each. FIG. 3-9 A shows a single endedversion. This can be an excellent topology for low cost concerns. FIG.3-9 B shows the effect of a transformer with a center tap. This circuitcan be useful but may not utilize the transformer secondary fully. Inaddition for low voltages some realizations can require the secondary tohave only one turn possibly making a center tap more difficult toimplement. FIG. 3-9 C shows inverting the SR's and the filter inductors.This circuit can be almost identical to the preferred one. In addition,the gate drive may not be referenced to a common source point making thedrive circuit more complex (not shown). FIG. 3-9 D shows a center tappedcoil in place of a center tapped secondary. Some magnetic realizationsmake this circuit attractive. The essentials of this disclosure apply aswell.

The above examples represent only a few of the many designs possible. Itshould be obvious from these variations that other circuits may bedesigned which embody the ideas disclosed.

Third Harmonic Trap

A series connection of an inductor and capacitor tuned to the thirdharmonic can be placed across the primary of the main VRM transformer.The preferred embodiment disclosed can draw an input current withsubstantial third harmonic content. By placing a trap on the input ofthe circuit the harmonic currents can flow through the trap and may notappear on the distribution supplying the circuit.

More importantly, the efficiency of the rectifier can be improved withthe addition of a third harmonic trap. The output circuit can benon-linear especially with the SR's having a long conduction angle (seeFIG. 3-4).

The DC output voltage from this circuit (FIG. 3-4 & 3-10) can be equalto the integral of the voltage across the SR's (the average voltageacross an inductor must be zero). Any distortion of this waveform canusually cause a reduction of the DC output voltage and consequently areduction in efficiency. The third harmonic trap can preserve thenatural peak of the SR voltage waveform.

Another potential benefit of the third harmonic trap is improvedstability of a system where multiple SR circuits are fed from a commonAC source. A local third harmonic trap can prevent SR circuits frominteracting due to third harmonic current flowing along the distributionpath.

Better put, without a third harmonic trap negative impedance can existduring a SR non-conduction time. Slight phase variations between SRcircuits can result in high harmonic energy flowing between SR circuits.This can manifest itself in overall system instability. The presence ofa third harmonic trap on the input of each SR circuit can locallysatisfy the high order current requirement and can result in systemstability.

Remote Power

Devices like microprocessors can require low voltage, high current andexhibit high di/dt requirements. In the circuit of FIG. 3-10, oneproblem which can exist is the di/dt limitation caused by theinterconnect inductance 326. In this commonly used circuit, bypasscapacitor 328 (which may be composed by many small capacitors inparallel) can be located near the microprocessor power pins. A largercapacitor, often called the bulk capacitor 327, can be located a smalldistance away. The short distance between capacitors 327 and 328 canform an inductor 326. This inductor 326 may limit the maximum di/dt themicroprocessor can pull from the power supply. This can be especiallytrue if the bypass capacitor is small (this is normally the case) and/orthe basic power conversion frequency is too low (also the normal case).The bypass capacitor 328 may not be kept charged to the demandedvoltage. Even if the power supply feeding capacitor 327 were ideal, orif capacitor 327 were replaced with an ideal voltage source a di/dtlimit might still exist as a result of the interconnect inductance 326.

In the circuit of the invention this problem can be overcome. Referringto FIG. 3-3, with this method and circuit the power conversion frequencycan be increased to the point where the output capacitance can be smallenough to be used as the microprocessor bypass capacitor which can belocated adjacent to the microprocessor power pins.

Quiet Power

One of the problems facing the power supply industry as voltages drop,currents increase and i/dt requirements increase is noise. The circuitof FIG. 3-1 is noisy for three reasons.

First the switching FET's 329 may be force commutated with steep voltagewavefronts. This can conduct and radiate noise into the surroundingstructures. Compare the voltage waveforms of FIG. 3-2 to those of FIG.3-4 to see the difference.

Second, the input circuit shown in FIG. 3-1 can inject current into theground path. As the FET's 329 are switched, large current can flowaround loop 330 through the input capacitor 332, interconnect inductance331 and FET's 329. The rate of change of current di/dt around this loop330 can cause a voltage to be developed across inductor 331 which can beimpressed onto the output voltage.

Third, the output of a circuit like FIG. 3-1 can be inherently noisy asthe DC output voltage is reduced. The DC output voltage is the averagevalue of the voltage on point 2 shown in FIG. 3-2. The voltageregulation method is sometimes dubbed pulse width modulation. For loweroutput voltages the pulse width becomes narrower to the point ofdifficulty of control. This is because a variation in width is a largerpercentage of the total pulse width. This can create a shaky or noisyoutput voltage.

The circuits being disclosed can use zero voltage switching (ZVS) andcan have smooth voltage waveforms in the rectification circuitry.Compare the voltage waveforms for FIG. 32 (Prior Art) to FIG. 3-4. It isobvious the waveforms on the invention will be less noisy. Secondly inone preferred embodiment the regulation can occur on the primary of thetransformer. This circuit is also ZVS plus it is isolated from the DCoutput voltage. These factors combined can make this approach much moresuited to the next generation low voltage devices.

ADDITIONAL EXAMPLE

FIGS. 3-11 and 3-12 show schematics for a complete ac to dc powerconverter which can include the rectifier section, the gate drive, theseries switch(es) along with a self derived dc power supply and feedbackfrom the output to the series switch for regulation. These schematicscan embody much of what has been disclosed and can show a completeworking 1.8 volt, 20 ampere DC power supply suitable for loads requiringhigh di/dt. They can operate from an AC input buss at 30 volts RMS at afrequency of 3.39 MHz. Finally, FIG. 3-13 shows a potential design forsome significant overall portions of the “silver box” as it may beconfigured in one preferred design.

The discussion included in this patent is intended to serve as a basicdescription. The reader should be aware that the specific discussion maynot explicitly describe all embodiments possible; many alternatives areimplicit. It also may not fully explain the generic nature of theinvention and may not explicitly show how each feature or element canactually be representative of a broader function or of a great varietyof alternative or equivalent elements. Again, these are implicitlyincluded in this disclosure. Where the invention is described indevice-oriented terminology, each element of the device implicitlyperforms a function. Apparatus claims may not only be included for thedevice described, but also method or process claims may be included toaddress the functions the invention and each element performs, andproduct by process claims may be added to address any system which uses,any product made through, and any item resulting from the inventiondisclosed herein. Neither the description nor the terminology isintended to limit the scope of the patent disclosure. It should also beunderstood that a variety of changes may be made without departing fromthe essence of the invention. Such changes are also implicitly includedin the description. They still fall within the scope of this invention.

The use of the principles described heretofore may result in a widevariety of configurations, and as above mentioned, this may permit awide variety of design tradeoffs. That is to say, this invention can beembodied in a wide variety of ways. In addition, each of the variouselements of the invention and claims may also be achieved in a varietyof manners. This disclosure should be understood to encompass each suchvariation, be it a variation of an embodiment of any apparatusembodiment, a method or process embodiment, or even merely a variationof any element of these. Particularly, it should be understood that asthe disclosure relates to elements of the invention, the words for eachelement may be expressed by equivalent apparatus terms or methodterms—even if only the function or result is the same. Such equivalent,broader, or even more generic terms should be considered to beencompassed in the description of each element or action. Such terms canbe substituted where desired to make explicit the implicitly broadcoverage to which this invention is entitled. As but one example, itshould be understood that all action may be expressed as a means fortaking that action or as an element which causes that action. Similarly,each physical element disclosed should be understood to encompass adisclosure of the action which that physical element facilitates.Regarding this last aspect, the disclosure of a “switch” should beunderstood to encompass disclosure of the act of “switching”—whetherexplicitly discussed or not—and, conversely, were there only disclosureof the act of “switching”, such a disclosure should be understood toencompass disclosure of a “switch.” Such changes and alternative termsare to be understood to be explicitly included in the description.

The foregoing discussion and the claims which follow describe thepreferred embodiments of the invention. Particularly with respect to theclaims it should be understood that changes may be made withoutdeparting from their essence. In this regard it is intended that suchchanges would still fall within the scope of the present invention. Itis simply not practical to describe and claim all possible revisionswhich may be accomplished to the present invention. To the extent suchrevisions utilize the essence of the invention each would naturally fallwithin the breadth of protection accomplished by this patent. This isparticularly true for the present invention since its basic concepts andunderstandings are fundamental in nature and can be applied in a varietyof ways to a variety of fields.

In addition, it should be understood that although claims directed tothe apparatus have been included in various detail, for administrativeefficiencies, only initial claims directed toward the methods have beenincluded. Naturally, the disclosure and claiming of the apparatus focusin detail is to be understood as sufficient to support the full scope ofboth method and apparatus claims. Additional method claims may be addedat a later date when appropriate to explicitly claim such details. Thus,the present disclosure is to be construed as encompassing the full scopeof method claims, including but not limited to claims and subclaimssimilar to those presented in a apparatus context. In addition otherclaims for embodiments disclosed but not yet claimed may be added aswell.

Furthermore, any references mentioned in the application for this patentas well as all references listed in any list of references filed withthe application are hereby incorporated by reference, however, to theextent statements might be considered inconsistent with the patenting ofthis invention such statements are expressly not to be considered asmade by the applicant(s).

Embodiments should be understood as encompassing both general powerdevices and tailored devices to specify a field or system to which it isapplied (such as a computer power supply in a computer system, or acomputer system having such a power supply, or the like), or may bepresented in combination with other elements. The various techniques anddevices disclosed represent a portion of that which those skilled in theart would readily understand from the teachings of this application.Additionally, the various combinations and permutations of all elementsor applications can be created and presented. All can be done tooptimize performance in a specific application. Further, unless thecontext requires otherwise, the word “comprise” or variations such as“comprises” or “comprising”, should be understood to imply the inclusionof a stated element or step or group of elements or steps but not theexclusion of any other element or step or group of elements or steps.

What is claimed is:
 1. A radio frequency power generator to providepower to a load comprising: a. a supply of power; b. a substantiallysinusoidal AC drive element with a drive amplitude; c. multiple switchesresponsive to said AC drive element and said supply of power whereinsaid multiple switches establish a alternating power output at afrequency; d. direct drive bias alteration circuitry to which said ACdrive element is responsive and which is responsive to said driveamplitude; and e. a lead which is responsive to said alternating poweroutput.
 2. A radio frequency power generator as described in claim 1wherein said multiple switches comprise two switches which operate inconjunction to establish said alternating power output.
 3. A radiofrequency power generator as described in claim 2 wherein said twoswitches comprise a half bridge configuration.
 4. A radio frequencypower generator as described in claim 3 wherein said two switches aresequentially operated and act to create transition events in betweeneither switch being in a conductive state, wherein each of saidtransition events has a transition period, and wherein said directcontrol network controls said drive bias alteration circuitry tomaintain the duration of said transition period.
 5. A radio frequencypower generator as described in claim 3 wherein said drive biasalteration circuitry comprises a drive bias alteration circuit for eachof said switches, and wherein said direct control network comprises adirect control network for each of said switches.
 6. A radio frequencypower generator as described in claim 4 wherein said two switches eachestablish a conduction angle, and wherein said direct control networkmaintains said conduction angles.
 7. A radio frequency power generatoras described in claim 1, wherein said direct drive bias alterationcircuitry comprises: a. voltage divider circuitry; and b. at least onediode element.
 8. A radio frequency power generator as described inclaim 6 and further comprising constant output voltage circuitry whichis responsive to said alternating power output of said multipleswitches.
 9. A radio frequency power generator as described in claim 8and further comprising constant trajectory circuitry which is alsoresponsive to said alternating power output of said multiple switches.10. A radio frequency power generator as described in claim 9 andfurther comprising energy maintenance circuitry which is also responsiveto said alternating power output of said multiple switches.
 11. A radiofrequency power generator as described in claim 10 and furthercomprising stabilizing circuitry which is also responsive to saidalternating power output of said multiple switches.
 12. A radiofrequency power generator as described in claim 8, 9, 10, or 11 whereinsaid multiple switches have output capacitance, and wherein saidcircuitry is tuned to coordinate with the frequency of said alternatingpower output and output capacitance.
 13. A radio frequency powergenerator as described in claim 12 wherein said substantially sinusoidalAC drive element comprises a high frequency driver, and wherein saidmultiple switches establish a high frequency alternating power output.14. A radio frequency power generator as described in claim 13 whereinsaid direct control network comprises a network with no feedback system.15. A system as described in claim 1 wherein said frequency drivercomprises a frequency driver operating at a frequency selected from agroup consisting of: a frequency greater than at least about 300 kHz, afrequency greater than at least about 500 kHz, a frequency greater thanat least about 1 MHz, a frequency greater than at least about 3 MHz, afrequency greater than at least about 10 MHz, and a frequency greaterthan at least about 30 MHz.
 16. A system as described in claim 1 whereinsaid variable load is capable of a rapid current demand which rises at alevel selected from a group consisting of: at least about 0.2 amperesper nanosecond, at least about 0.5 amperes per nanosecond, at leastabout 1 ampere per nanosecond, at least about 3 amperes per nanosecond,at least about 10 amperes per nanosecond, and at least about 30 amperesper nanosecond.
 17. A system as described in claim 1 wherein saidnetwork comprises a fast acting response network.
 18. A system asdescribed in claim 17 wherein said fast acting response networkcomprises a network having an effective capacitance selected from agroup consisting of: less than about 0.3 millifarads, less than about0.5 millifarads, less than about 1 millifarads, less than about 3millifarads, less than about 10 millifarads.
 19. A system as describedin claim 17 wherein said fast acting response network comprises aresponse network which is capable of reacting within a period of timeselected from a group consisting of: less than about a period of aNyquist frequency, less than about two and a half times a period of aNyquist frequency, less than about five times a period of a Nyquistfrequency, less than about ten times a period of a Nyquist frequency,less than about twice a period of said alternating power output, lessthan about four times a period of said alternating power output, lessthan about 200 nanoseconds, less than about 500 nanoseconds, less thanabout 1000 nanoseconds, and less than about 2000 nanoseconds.
 20. Asystem as described in claim 1 wherein said variable load comprises aload operating at a nominal DC voltage selected from a group consistingof: less than about 2 volts, less then about 1.8 volts, less than about1.5 volts, less than about 1.3 volts, less than about 1 volt, and lessthan about 0.4 volts.
 21. A system as described in claim 20 wherein saidload comprises a load operating at a maximum current selected from agroup consisting of: more than about 15 amperes, more than about 20amperes, and more than about 50 amperes.
 22. A system as described inclaim 1 wherein said network comprises a high efficiency responsenetwork.
 23. A system as described in claim 22 wherein said highefficiency response network comprises a response network having anefficiency selected from a group consisting of: at least about 80%, atleast about 85%, at least about 90%, at least about 95%, at least about98% and at least about 99%.
 24. A system as described in claim 17wherein said network comprises a network with no feedback system.
 25. Amethod of generating radio frequency power to provide power to a loadcomprising the steps of: a. supplying power; b. inverting said powerthrough multiple switches to establish an alternating power output; c.substantially sinusoidally driving said switches with a drive amplitudeand utilizing a drive bias to create an alternating power output; d.directly altering said drive bias through circuitry which is responsiveto said drive amplitude; and e. powering a variable load responsive tosaid alternating power output.
 26. A method of generating radiofrequency power to provide power to a load as described in claim 25wherein said step of inverting said power through multiple switches toestablish an alternating power output comprises the step of sequentiallyoperating two switches to create transition events in between eitherswitch being in a conductive state in a manner which is responsive tosaid step of directly controlling said drive bias through a network towhich said drive bias is responsive.
 27. A method of generating radiofrequency power to provide power to a load as described in claim 26wherein each switch has a drive bias, and wherein said step ofsubstantially sinusoidally driving said switches utilizing a drive biasto create an alternating power output comprises the step of directlycontrolling the drive bias of each switch.
 28. A method of generatingradio frequency power to provide power to a load as described in claim25 and further comprising the steps of: a. establishing a constantoutput voltage through circuitry which is responsive to said alternatingpower output of said switch; b. establishing a constant trajectorythrough circuitry which is also responsive to said alternating poweroutput of said switch; c. maintaining component supply energy throughcircuitry which is also responsive to said alternating power output ofsaid switch; and d. stabilizing said alternating power output throughcircuitry which is also responsive to said alternating power output ofsaid switch.
 29. A method of generating radio frequency power to providepower to a load as described in claim 28 wherein said switches haveoutput capacitance, and further comprising the step of tuning saidcircuitry to said frequency of operation and said output capacitance.30. A method of generating radio frequency power to provide power to aload as described in claim 29 wherein said step of substantiallysinusoidally driving said switches utilizing a drive bias to create analternating power output comprises the step of high frequency drivingsaid switch to create a high frequency alternating power output.
 31. Ahigh frequency power generator to provide power to a load comprising: a.a supply of power; b. a high frequency driver; c. at least one switchresponsive to said high frequency driver and said supply of power,having a body diode feature, wherein said at least one switchestablishes a high frequency alternating power output; d. a variableload which is responsive to said high frequency alternating poweroutput; and e. a passive conduction prevention response network which isresponsive to said high frequency alternating power output of saidswitch and which prevents said body diode feature from transitioning toa conduction state, said network comprising: i. constant output voltagecircuitry; ii. constant trajectory circuitry; iii. a drive biasalteration circuitry to which said driver is responsive and having acontrol input; and iv. a direct control network which provides thecontrol input to said drive bias alteration circuitry.